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DataSheet.in
LT3748 100V Isolated Flyback Controller Features
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Description
The LT(R)3748 is a switching regulator controller specifically designed for the isolated flyback topology and capable of high power. It drives a low side external N-channel power MOSFET from an internally regulated 7V supply. No third winding or opto-isolator is required for regulation as the part senses the isolated output voltage directly from the primary-side flyback waveform. The LT3748 utilizes boundary mode to provide a small magnetic solution without compromising load regulation. Operating frequency is set by load current and transformer magnetizing inductance. The gate drive of the LT3748 combined with a suitable external MOSFET allow it to deliver load power up to several tens of watts from input voltages as high as 100V. The LT3748 is available in a high voltage 16-lead MSOP package with four leads removed.
L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S. Patents, including 5438499 and 7471522.
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5V to 100V Input Voltage Range 1.9A Average Gate Drive Source and Sink Current Boundary Mode Operation No Transformer Third Winding or Opto-Isolator Required for Regulation Primary-Side Winding Feedback Load Regulation VOUT Set with Two External Resistors INTVCC Pin for Control of Gate Driver Voltage Programmable Soft Start Programmable Undervoltage Lockout Available in MSOP Package
applications
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Isolated Telecom Converters High Power Automotive Supplies Isolated Industrial Power Supplies
typical application
25W, 12V Output, Isolated Telecom Supply
VIN 36V TO 72V 10F 412k EN/UVLO 15.4k VIN RFB RREF LT3748 TC SS VC 56.2k 2nF 10k 4700pF 4.7F GND GATE SENSE INTVCC 0.033
3748 TA01a
4:1 60.8H 243k 3.8H
VOUT+ 12V 2A 100F VOUT-
Output Load and Line Regulation
12.6 14.4 12.2 VOUT (V) 12.0 11.8 11.6 11.4 VIN = 72V VIN = 48V VIN = 36V 0 0.5 1.0 1.5 LOAD CURRENT (A) 2.0
3748 TA01b
6.04k
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DataSheet.in
LT3748 absolute MaxiMuM ratings
(Note 1)
pin conFiguration
TOP VIEW VIN 1 EN/UVLO 3 INTVCC GATE SENSE GND 5 6 7 8 16 RFB 14 RREF 12 11 10 9 TC VC SS GND
VIN, RFB ...................................................................100V VIN to RFB..................................................................5V EN/UVLO......................................................-0.3V, 100V INTVCC ....................................................VIN + 0.3V, 20V SS, VC, TC, RREF .........................................................6V SENSE ......................................................................0.4V Maximum Junction Temperature .......................... 125C Operating Junction Temperature Range (Note 2).................................................. -40C to 125C Storage Temperature Range .................. -65C to 150C
MS PACKAGE 16 (12)-LEAD PLASTIC MSOP TJMAX = 125C, JA = 90C/W
orDer inForMation
LEAD FREE FINISH LT3748EMS#PBF LT3748IMS#PBF TAPE AND REEL LT3748EMS#TRPBF LT3748IMS#TRPBF PART MARKING* 3748 3748 PACKAGE DESCRIPTION 16-Lead Plastic MSOP 16-Lead Plastic MSOP TEMPERATURE RANGE -40C to 125C -40C to 125C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
electrical characteristics
PARAMETER Input Voltage Range Quiescent Current VIN Quiescent Current, INTVCC Overdriven INTVCC Voltage Range INTVCC Pin Regulation Voltage INTVCC Dropout INTVCC Undervoltage Lockout EN/UVLO Pin Threshold EN/UVLO Pin Hysteresis Current Soft-Start Current Soft-Start Threshold Soft-Start Reset Current Maximum SENSE Current Limit Threshold Minimum SENSE Current Limit Threshold Maximum to Minimum SENSE Threshold Ratio VC = 2.2V Not Switching VEN/UVLO = 0.2V VINTVCC = 10V CONDITIONS
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 10V, unless otherwise noted.
MIN
l
TYP 1.3 0 300
MAX 100 1.75 1 450 20 7.2 3.75 1.25 2.9
UNITS V mA A A V V V V V A A V mA
5
l
4.5 6.8 7 0.7 3.6 1.223 2.4 5 0.65 3 95 90 5.2 100 100 15 6.6
(VIN - VINTVCC), IINTVCC = 10mA, VIN = 5V Falling Threshold EN/UVLO Pin Voltage Rising EN/UVLO = 1V VSS = 0.4V (Note 3)
l l
3.45 1.19 1.9
l
105 110 8.2
mV mV mV mV/mV
VC = 0V
l
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LT3748 electrical characteristics
PARAMETER SENSE Overcurrent Threshold SENSE Input Bias Current RREF Voltage RREF Voltage Line Regulation RREF Pin Bias Current TC Current into RREF Error Amplifier Voltage Gain Error Amplifier Transconductance VC Source Current VC Sink Current Flyback Comparator Trip Current Minimum GATE Off-Time Minimum GATE On-Time Maximum Discontinuous Off-Time Maximum GATE Off-Time Maximum GATE On-Time GATE Output Rise Time GATE Output Fall Time GATE Output Low (VOL) GATE Output High (VOH) Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LT3748E is guaranteed to meet performance specifications from 0C to 125C junction temperature. Specifications over the -40C VINTVCC - 0.05 to 125C operating junction temperature range are assured by design characterization and correlation with statistical process controls. The LT3748I is guaranteed over the full -40C to 125C operating junction temperature range. Note 3: Current flows out of the pin. VC = 0V VRREF = 0.5V VSENSE = 0V , CL = 3300pF 10% to 90% , CL = 3300pF 10% to 90% I = 10A VC = 1.1V, VRREF = 0.5V VC = 1.1V, VRREF = 2V Current into RFB Pin, RREF = 6.04k CONDITIONS VC = 2.2V VSENSE = 10mV (Note 3) VC = 1.1V
l
The l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25C. VIN = 10V, unless otherwise noted.
MIN 115 10 1.20 1.195 TYP 130 15 1.223 0.005
l
MAX 145 20 1.24 1.245 0.025 500
UNITS mV A V V %/V nA A V/V mhos A A A ns ns s s s ns ns
5V < VIN < 100V (Note 3) RTC = 20k
35 27.5 115 155 -45 48 10 700 250 24 55 55 16 16
0.05
V V
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DataSheet.in
LT3748 typical perForMance characteristics TA = 25C, unless otherwise noted.
Output Regulation vs Temperature
15.6 15.4 15.2 VOUT (V) 15.0 14.8 14.6 14.4 -50 -25 FIGURE 15 CIRCUIT IOUT = 150mA ON EACH OUTPUT VIN = 12V 1.7
Quiescent Current vs Temperature
VSS = 0V 1.6 INTVCC = OPEN QUIESCENT CURRENT (mA) QUIESCENT CURRENT (mA) 1.5 1.4 1.3 1.2 1.1 1.0 0.9 VIN = 12V VIN = 6V VIN = 72V VIN = 36V 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 25 50 75 100 125 150 TEMPERATURE (C)
3748 G02
Quiescent Current vs VIN Voltage
VSS = 0V INTVCC = OPEN
0
25 50 75 100 125 150 TEMPERATURE ( C)
3748 G01
0.8 -50 -25
0
0
20
40
60 VIN (V)
80
100
3748 G03
7.5 7.4 7.3 INTVCC VOLTAGE (V) 7.2
INTVCC Voltage vs Temperature
7.5 7.0 6.5 VINTVCC (V) IINTVCC = 0mA 6.0 5.5 5.0 4.5 0 25 50 75 100 125 150 TEMPERATURE (C)
3748 G04
INTVCC Voltage vs VIN Voltage
IINTVCC = 0mA IINTVCC = 10mA INTVCC UVLO (V)
INTVCC Undervoltage Lockout vs Temperature
4.0 3.9 3.8 3.7 3.6 3.5 3.4 RISING THRESHOLD FALLING THRESHOLD
7.1 7.0 6.9 6.8 6.7 6.6 6.5 -50 -25
IINTVCC = 10mA
4.0
4
6
8
10
20
40
60
80
100
3.3 -50 -25
0
VIN VOLTAGE (V)
3748 G05
25 50 75 100 125 150 TEMPERATURE (C)
3748 G06
3.0 INTVCC REGULATOR DROPOUT (V) 2.5
INTVCC Regulator Dropout vs INTVCC Current
VIN = 5V 3.0 2.5 INTVCC DROPOUT (V) 2.0 1.5
INTVCC Dropout vs Temperature
VIN = 5V SOFT-START CURRENT (A) 6 5 4 3 2 1
Soft-Start Current vs Temperature
2.0 1.5 1.0 0.5 0 150C 100C 25C -50C 0 10 20 30 INTVCC CURRENT (mA) 40
3748 G07
IINTVCC = 20mA IINTVCC = 10mA
1.0 0.5 IINTVCC = 5mA 0 -50 -25 0 25 50 75 100 125 150 TEMPERATURE ( C)
3748 G08
0 -50 -25
0
25 50 75 100 125 150 TEMPERATURE ( C)
3748 G09
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DataSheet.in
LT3748 typical perForMance characteristics
EN/UVLO Current vs Temperature
3.0 2.5 EN/UVLO CURRENT (A) 2.0 1.5 1.0 0.5 0 -50 -25 VEN/UVLO = 1.3V 0 25 50 75 100 125 150 TEMPERATURE (C)
3748 G10
TA = 25C, unless otherwise noted.
EN/UVLO Threshold vs Temperature
1.40 1.35 EN/UVLO THRESHOLD (V) 0.9 0.8 0.7 TC VOLTAGE (V) 0.6 0.5 0.4 0.3 0.2 0.1 0 25 50 75 100 125 150 TEMPERATURE (C)
3748 G11
TC Pin Voltage vs Temperature
VEN/UVLO = 1.1V VEN/UVLO = 0.9V
1.30 1.25 1.20 1.15 1.10 1.05 1.00 -50 -25
0 -50 -25
0
25 50 75 100 125 150 TEMPERATURE (C)
3748 G12
Error Amplifier Transconductance vs Temperature
200 190 TRANSCONDUCTANCE (mhos) 180 170 150 140 130 120 110 100 -50 -25 0 VIN = 100V VIN = 6V 25 50 75 100 125 150 TEMPERATURE (C)
3748 G13
60 50 40 30 20 IVC (A) 10 0 -10 -20 -30 -40 -50 -60
Error Amplifier Output Current vs RREF Pin Voltage
160 140 SENSE THRESHOLD (mV) 120 100 80 60 40 20 1.5 1.0 VREF (V) 2.0 2.5
3748 G14
SENSE Pin Threshold vs Temperature
OVERCURRENT
160
VC = 2.2V
150C 100C 25C -50C 0 0.5
VC = 0.2V 0 25 50 75 100 125 150 TEMPERATURE (C)
3748 G15
0 -50 -25
Maximum Discontinuous Off-Time vs Temperature
30 MAXIMUM DISCONTINUOUS OFF-TIME (s) 29 GATE RISE AND FALL TIME (ns) 28 27 26 25 24 23 22 21 20 -50 -25 0 25 50 75 100 125 150 TEMPERATURE (C)
3748 G16
GATE Rise and Fall Time vs Charge
2.0 AVERAGE CURRENT 50 40 RISE TIME 30 FALL TIME 20 10 0 0 20 Q=C*V VINTVCC = 7V tr, tf 10% TO 90% 40 60 80 100 TOTAL GATE CHARGE (nC) 120 0 1.5 1.0 0.5 25 GATE RISE AND FALL TIME (ns) 20 AVERAGE GATE SOURCE, SINK CURRENT (A)
GATE Rise and Fall Time vs INTVCC Voltage
CGATE = 3.3nF tr, tf 10% TO 90% FALLING RISING
15 10
5
0
0
5
10 VINTVCC (V)
15
20
3748 G18
3748 G17
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DataSheet.in
LT3748 pin Functions
VIN (Pin 1) Input Voltage. This pin supplies current to the internal start-up circuitry and is the reference voltage for the feedback circuitry connected to the RFB pin. This pin must be locally bypassed with a capacitor. EN/UVLO (Pin 3): Enable/Undervoltage Lockout. A resistor divider connected to VIN is tied to this pin to program the minimum input voltage at which the LT3748 will operate. At a voltage below ~0.5V, the part draws less than 1A quiescent current. When below 1.223V but above ~0.5V, the part will draw quiescent current but will not regulate the INTVCC supply or power the gate drive circuitry. Above 1.223V, all internal circuitry will start and the SS pin will source 5A. When EN/UVLO falls below 1.223V, 2.4A is sunk from the pin to provide programmable hysteresis for undervoltage lockout. INTVCC (Pin 5): Gate Driver Bias Voltage. This pin supplies current to the internal gate driver circuitry of the LT3748. The INTVCC pin must be locally bypassed with a capacitor. This pin may also be connected to VIN if a third winding is not used and if VIN 20V. If a third winding is used, the INTVCC voltage should be lower than the input voltage for proper operation. GATE (Pin 6): N-Channel MOSFET Gate Driver Output. Switches between INTVCC and GND. SENSE (Pin 7): The Current Sense Input for the Control Loop. Kelvin connect this pin to the positive terminal of the switch current sense resistor, RSENSE, in the source of the N-channel MOSFET. The negative terminal of the current sense resistor should be connected to the GND plane close to the IC. GND (Pins 8, 9): Ground. SS (Pin 10): Soft-Start Pin. This pin delays start-up and clamps VC pin voltage. Soft-start timing is set by the size of the external capacitor at the pin. Switching starts when VSS reaches ~0.65V. VC (Pin 11): Compensation Pin for the Internal Error Amplifier. Connect a series RC from this pin to ground to compensate the switching regulator. A 100pF capacitor in parallel helps eliminate noise. TC (Pin 12): Output Voltage Temperature Compensation. Connect a resistor to ground to produce a current proportional to absolute temperature to be sourced into the RREF node. ITC = 0.55V/RTC. RREF (Pin 14): Input Pin for the External Ground-Referred Reference Resistor. The resistor at this pin should be 6.04k, but for convenience in selecting a resistor divider ratio, the value may range from 5.76k to 6.34k. The resistor should be as close to the LT3748 as possible. RFB (Pin 16): Input Pin for the External Feedback Resistor. This pin is connected to the transformer primary at the external MOSFET power switch. The ratio of this resistor to the RREF resistor, times the internal bandgap reference, determines the output voltage (plus the effect of any non-unity transformer turns ratio). The average current through this resistor during the flyback period should be approximately 200A. The resistor should be as close to the LT3748 as possible.
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DataSheet.in
LT3748 block DiagraM
VIN CIN RFB
1 16
T1 DOUT NPS:1 LPRI LSEC
VOUT + COUT VOUT -
TC CURRENT Q1
12
VIN
RFB Q2 BOUNDARY MODE DETECT 1.223V
TC
RTC 20A 6.04k
-A1 +
ERROR AMP 1.223V
+ A4 -
INTVCC
5
CBIAS 50s MAX OFF TIMER MASTER LATCH S Q R
14
RREF
+ gm -
RREF VARIABLE DELAY TIMER S R
A4
GATE
6
NMOS
R1
3
1.223V EN/UVLO
+ A3 -
R2 2.4A
INTERNAL REFERENCE AND REGULATORS
5A
50s MAX ON TIMER
GND 8, 9
--+
100mV CURRENT LIMIT
A2
SENSE
7
RSENSE VC
11
10
SS
CSS
3748 BD
RC CC
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DataSheet.in
LT3748 operation
The LT3748 is a current mode switching regulator controller designed specifically for the isolated flyback topology. The special problem normally encountered in such circuits is that information relating to the output voltage on the isolated secondary side of the transformer must be communicated to the primary side in order to maintain regulation. Historically, this has been done with optoisolators or extra transformer windings. Opto-isolator circuits waste output power and the extra components increase the cost and physical size of the power supply. Opto-isolators can also exhibit trouble due to limited dynamic response, nonlinearity, unit-to-unit variation and aging over life. Circuits employing extra transformer windings also exhibit deficiencies. Using an extra winding adds to the transformer's physical size and cost, and dynamic response is often mediocre. The LT3748 derives its information about the isolated output voltage by examining the primary-side flyback pulse waveform. In this manner, no opto-isolator nor extra transformer winding is required for regulation. The output voltage is easily programmed with two resistors. The LT3748 features a boundary mode control method, (also called critical conduction mode) where the part operates at the boundary between continuous conduction mode and discontinuous conduction mode. Due to the boundary control mode operation, the output voltage can be calculated from the transformer primary voltage when the secondary current is almost zero. This method improves load regulation without external resistors and capacitors. The Block Diagram shows an overall view of the system. Many of the blocks are similar to those found in traditional switching regulators, including current comparators, internal reference and regulators, logic, timers and an N-channel MOSFET gate driver. The novel sections include a special sampling error amplifier and a temperature compensation circuit. Boundary Mode Operation Boundary mode is a variable frequency, current mode switching scheme. The external N-channel MOSFET turns on and the inductor current increases until it reaches the VC pin-controlled current limit. After the external MOSFET is turned off, the voltage on the drain of the MOSFET rises to the output voltage multiplied by the primary-to-secondary transformer turns ratio plus the input voltage. When the secondary current through the output diode falls to zero, the voltage on the drain of the MOSFET falls below VIN . A boundary mode detection comparator detects this event and turns the external MOSFET back on. Boundary mode returns the secondary current to zero every cycle, so the parasitic resistive voltage drops do not cause load regulation errors. Boundary mode also allows the use of a smaller transformer compared to continuous conduction mode and does not exhibit subharmonic oscillation. At low output currents the LT3748 delays turning on the external MOSFET and thus operates in discontinuous mode. Unlike traditional flyback converters, the external MOSFET has to turn on to update the output voltage information. Below 0.6V on the VC pin, the current comparator level decreases to its minimum value and a variable delay timer waits to reset before turning on the external MOSFET. With the addition of delay before turning the MOSFET back on, the part starts to operate in discontinuous mode. The average output current is able to decrease while still allowing a minimum off-time for the error amplifier sampling circuitry. The typical maximum discontinuous off-time with VC equal to 0V is 24s.
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LT3748 applications inForMation
Pseudo-DC Theory of Operation The RREF and RFB resistors as depicted in the Block Diagram are external resistors used to program the output voltage. The LT3748 operates much the same way as traditional current mode switchers with the exception of the unique error amplifier which derives its feedback information from the flyback pulse. Operation is as follows: when the NMOS output switch turns off, its drain voltage rises above VIN. The amplitude of this flyback pulse (i.e., the difference between it and VIN) is given as: VFLBK = (VOUT + VF + ISEC * ESR) * NPS VF = DOUT forward voltage ISEC = Transformer secondary current ESR = Total impedance of secondary circuit NPS = Transformer effective primary-to-secondary turns ratio The flyback voltage is converted to a current by RFB and Q2. Nearly all of this current flows through resistor RREF to form a ground-referred voltage. This voltage is fed into the flyback error amplifier. The flyback error amplifier samples this output voltage information when the secondary-side winding current reaches zero. The error amplifier uses a bandgap voltage, 1.223V, as the reference voltage. The relatively high gain in the overall loop will then cause the voltage at the RREF resistor to be nearly equal to the bandgap reference voltage, VBG. The relationship between VFLBK and VBG may then be expressed as: VFLBK VBG or R =R FB REF R VFLBK = VBG FB RREF VBG = Internal bandgap reference Combining with the previous VFLBK expression yields an expression for VOUT, in terms of the internal reference, programming resistors, transformer turns ratio and diode forward voltage drop: R 1 VOUT = VBG FB - VF - ISEC (ESR) RREF NPS Additionally, it includes the effect of nonzero secondary output impedance (ESR). This term can be assumed to be zero in boundary control mode. Temperature Compensation The first term in the VOUT equation does not have a temperature dependence, but the diode forward drop, VF, has a significant negative temperature coefficient. To compensate for this, a positive temperature coefficient current source is internally connected to the RREF pin. The current is set by resistor RTC to ground connected between the TC pin and ground. To cancel the temperature coefficient, the following equation is used: d VF R 1 = - FB * * dT R TC NPS -RFB 1 R TC = * NPS d VF / d T d VTC or, dT dV R * TC FB dT NPS
(dVF /dT) = Diode's forward voltage temperature coefficient (dVTC /dT) = 1.85mV/C VTC = 0.55V The resistor value given by this equation should also be verified experimentally and adjusted, if necessary, to achieve optimal regulation over temperature. The revised output voltage is as follows: R 1 VOUT = VBG FB - VF RREF NPS V R - TC * FB - ISEC (ESR) R TC NPS
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DataSheet.in
LT3748 applications inForMation
Selecting Actual RREF , RFB and RTC Resistor Values The preceding equations define how the LT3748 would regulate the output voltage if the system had no time delays and no error sources. However, there are a number of repeatable delays and parasitics in each application which will affect the output voltage and force a re-evaluation of the RFB and RTC component values. The following approach is the best method for selecting the correct values. The expression for VOUT, developed in the Operation section, can be rearranged to yield the following expression for RFB: RFB = where: VOUT = Output voltage VF = Output diode forward voltage NPS = Effective primary-to-secondary turns ratio VTC = 0.55V The equation assumes the temperature coefficients of the output diode and VTC are equal and substitutes RFB/NPS for the value of RTC. This is a good first order approximation but will be revisited later. First, the value of RREF should be approximately 6.04k since the LT3748 is trimmed and specified using this value. If the impedance of RREF varies considerably from 6.04k, additional errors will result. However, a variation in RREF of several percent is acceptable. This yields a bit of freedom in selecting standard 1% resistor values to yield nominal RFB /RREF ratios. With starting values for RFB and RTC, an initial iteration of the application should be built with final selections of all external components (transformer, diode, MOSFET, etc.). The resulting VOUT should be measured and used to re-evaluate the value of RFB due to non-idealities in the sampling system: RFB(NEW ) = VOUT(DESIRED) VOUT(MEASURED) * RFB(OLD) RREF * NPS ( VOUT + VF ) + VTC VBG With a new value of RFB selected, the temperature coefficient of the output diode in the application can be tested to verify the nominal RTC value. The RTC resistor should be removed from the circuit under test (this will cause VOUT to increase for this step) and VOUT should be measured over temperature at a desired target output load. It is very important for this evaluation that uniform temperature be applied to both the output diode and the LT3748--if freeze spray or a heat gun is used there can be a significant mismatch in temperature between the two devices that causes significant error. Attempting to extrapolate the data from a diode datasheet or assuming the nominal RTC value may yield a better result if there is no method to apply uniform heat or cooling such as an oven. With at least two data points (although more data points from hot to cold are recommended), the change in V/C can be determined by: VOUT V -V = OUT1 OUT 2 TEMP TEMP1- TEMP2 Using the measured VOUT temperature coefficient, an exact RTC value can be selected using the following equation: R TC = RFB 1.85mV/C * VOUT NPS TEMP
If the value of RTC has changed significantly, which can happen with the use of some output diodes that have a very low forward drop, the RFB value may need to be changed to restore VOUT to the desired value. As in the previous iteration, after measuring VOUT , a new RFB can once again be selected using: RFB(NEW ) = VOUT(DESIRED) VOUT(MEASURED) * RFB(OLD)
Once the values of RFB and RTC are selected, the regulation accuracy from board to board for a given application will be very consistent, typically under 5% when including device variation of all the components in the system (assuming resistor tolerances and transformer windings matching of 1% or better). However, if the transformer, the output diode or MOSFET switch are changed or the layout is dramatically altered, there may be some change in VOUT .
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DataSheet.in
LT3748 applications inForMation
Minimum Primary Inductance Requirements The LT3748 obtains output voltage information from the external MOSFET drain voltage when the secondary winding conducts current. The sampling circuitry needs a minimum of 400ns to settle and sample the output voltage while the MOSFET switch is off. This required settle and sample time is controlled by external components independent of the minimum off-time of the GATE pin as specified in the Electrical Characteristics table. The electrical specification minimum off-time is based on an internal timer and acts as a maximum frequency clamp. The following equation gives the minimum value for primary-side magnetizing inductance:
L PRI
Output Power Because the MOSFET power switch is located outside the LT3748, the maximum output power is primarily limited by external components. Output power limitations can be separated into three categories--voltage limitations, current limitations and thermal limitations. The voltage limitations in a flyback design are primarily the MOSFET switch VDS(MAX) and the output diode reverse-bias rating. Increasing the voltage rating of either component will typically decrease application efficiency if all else is equal and the voltage requirements on each of those components will be directly related to the windings ratio of the transformer, the input and output voltages and the use of any additional snubbing components. The MOSFET VDS(MAX) must theoretically be higher than VIN(MAX) + (VOUT * NPS) and the output diode reverse bias must be higher than VOUT + (VIN(MAX)/NPS), though leakage inductance spikes on both the drain of the MOSFET and the anode of the output diode may more than double that requirement (see section on leakage inductance for more details on snubbers). Figure 1 illustrates the effect on available output power for several MOSFET voltage ratings while continuously maximizing windings ratio for input voltage with a fixed MOSFET current limit and output voltage. Increasing the MOSFET rating increases the possible windings ratio and or maximum input voltage and can increase the available output power for a given application. Both figures assume no leakage inductance and high efficiency.
50 VDS = 200V MAXIMUM OUTPUT POWER (W) 40 VDS = 150V
( VOUT + VF(DIODE) ) * RSENSE * t SETTLE(MIN) * NPS
VSE NSE(MIN)
VSENSE(MIN) = 15mV tSETTLE(MIN) = 400ns NPS = Ratio of primary windings to secondary windings In addition to the primary inductance requirement for minimum settling and sampling time, the LT3748 has internal circuit constraints that prevent it from setting the GATE node high for shorter than approximately 250ns. If the inductor current exceeds the desired current limit during that time oscillation may occur at the output as the current control loop will lose its ability to regulate. Therefore, the following equation relating to maximum input voltage must also be followed in selecting primaryside magnetizing inductance: LPRI VIN(MAX) *RSENSE * tON(MIN) VSENSE(MIN)
tON(MIN) = 250ns The last constraint on minimum inductance value would relate to minimum full-load operating frequency, fMIN, and is derived from fSW = 1/(tON + tOFF): LPRI VIN(MIN) * (VOUT + VF(DIODE)) * NPS/(fSW(MIN) * ILIM * ((VOUT + VF(DIODE)) * NPS + VIN(MIN))) The minimum operating frequency may be lower than the calculated number due to delays in detecting current limit and detecting boundary mode that are specific to each application.
30 VDS = 100V 20 10
0
0
20
60 40 INPUT VOLTAGE (V)
80
100
3748 F01
Figure 1. Maximum Output Power at 12VOUT with a 3A ILIM and Maximum VDS = 100V, 150V, 200V
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DataSheet.in
LT3748 applications inForMation
The current limitation on output power delivery is generally constrained by transformer saturation current in higher power applications, although the MOSFET switch and output diode will need to be rated for the desired currents, as well. Increasing the peak current on the primary side of the flyback by reducing the RSENSE resistor is the primary way to increase output power, and power delivered increases fairly linearly with current limit as shown in Figure 2, until parasitic losses begin to dominate. However, once the saturation current of the transformer is exceeded the energy coupling between the primary and the secondary will be reduced and incremental power will not be delivered to the output. In addition, the primary inductance will drop, the SENSE pin overcurrent threshold may trip due to a corresponding rapid rise in current, and the transformer will have to absorb the energy that is not transferred through the saturated core, leading to heating. Some manufacturers may not specify the rated saturation current but it is a necessary specification when trying to minimize transformer size and maximize output power and efficiency. Also necessary for proper design is data on saturation current over temperature--the saturation of typical power ferrites may reduce by over 20% from 25C to 100C. The thermal limitation in flyback applications for lower output voltages will be dominated by losses in the output diode, with resistive and leakage losses in the transformer
50 ILIM = 3A MAXIMUM OUTPUT POWER (W) 40 ILIM = 2A 30
increasing as a percentage basis of loss as the output voltage is increased. As power levels increase the output diode and transformer may exceed their rated temperature specifications. Minimizing RMS output diode current, selecting a diode with minimal forward drop at expected currents and minimizing parasitic resistances and leakage inductance in the transformer will keep those components below their maximum temperatures while maximizing efficiency. The following section discussing transformer selection will further help focus on how to minimize losses in the output diode. While quiescent current in the LT3748 itself is low (approximately 300A from VIN and 1mA from INTVCC), the current required to drive the external MOSFET (fSW * QG), if drawn from VIN through the LT3748 INTVCC LDO, dissipates (VIN - INTVCC) * fSW * QG. If that power is high enough to cause significant heating of the LT3748 the current may need to be drawn from a third winding. Doing so will push all thermal limitations outside of the LT3748. Selecting a Transformer Transformer specification and design is perhaps the most critical part of successfully applying the LT3748. In addition to the usual list of caveats dealing with high frequency isolated power supply transformer design, the following information should be carefully considered. First and most importantly, since the voltage on the secondary side of the transformer is inferred by the voltage sampled on the primary, the transformer turns ratio must be tightly controlled to ensure a consistent output voltage. A tolerance of 5% in turns ratio from transformer to transformer could result in a variation of more than 5% in output regulation. Fortunately, most magnetic component manufacturers are capable of guaranteeing a turns ratio tolerance of 1% or better. Linear Technology has worked with several leading magnetic component manufacturers to produce predesigned flyback transformers for use with the LT3748. Table 1 shows the details of several of these transformers.
20 10
ILIM = 1A
0
0
20
60 40 INPUT VOLTAGE (V)
80
100
3748 F02
Figure 2. Maximum Output Power at 12VOUT with 150V VDS(MAX) and ILIM = 1A, 2A, 3A
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Table 1. Pre-Designed Transformers--Typical Specifications Unless Otherwise Noted
TRANSFORMER PART NUMBER 750311424 750311456* 750311439 750311423 750311457 750311689 750311458* 750311564 750311624 750311604 750311599 750311600 750311608 750311607 750311590 750311591 750311592 750311594 750311595 750311596 PA2367NL PA1276NL PA2467NL PA1260NL PA3177NL Size (W x L x H) mm 17.7 x 14.0 x 12.7 17.7 x 14.0 x 12.7 17.7 x 14.0 x 12.7 17.7 x 14.0 x 12.7 17.7 x 14.0 x 12.7 17.7 x 14.0 x 12.7 17.7 x 14.0 x 12.7 17.7 x 14.0 x 12.7 17.7 x 14.0 x 12.7 29.08 x 23.11 x 11.43 29.08 x 23.11 x 11.43 29.08 x 23.11 x 11.43 29.08 x 23.11 x 11.43 29.08 x 23.11 x 11.43 32.31 x 27.03 x 13.69 32.31 x 27.03 x 13.69 32.31 x 27.03 x 13.69 32.31 x 27.03 x 13.69 32.31 x 27.03 x 13.69 32.31 x 27.03 x 13.69 17.7 x 14.0 x 12.7 17.7 x 14.0 x 12.7 17.7 x 14.0 x 12.7 17.7 x 14.0 x 12.7 29.21 x 21.84 x 11.43 LPRI (H) 100 100 37 50 50 50 15 9 9 8 8 12 12 14 8 8 8 15 12 12 85 77.4 37 77.4 8.3 LLEAK (nH) 844 900 750 570 600 600 175 120 150 300 500 500 500 500 200 200 200 400 200 200 750 800 750 800 100 NPS (NP:NS) 3:1 3:1 2:1 4:1 4:1 4:1 3:1 3:1 1.5:1 1:1 1.5:1 3:1 1.5:1 2.5:1 2:1 1.5:1 1:1 2.33:1 3:1 1.5:1 2.7:1 1.47:1 2:1 3.67:1 2:1 ISAT (A) 3 2.4 2.8 4 3.7 3.7 5 8 8 9.5 12 11 9 9.5 18 20 18 18 18 16 1.7 1.6 2.9 1.5 8.6 RPRI (m) 180 225 89 90 115 115 35 36 34 30 30 30 30 40 15 15 15 35 15 30 325 100 89 220 10 RSEC (m) 29 31 28 12 12 12 6 7 21 12 12 40 20 10 8 12 20 15 12 30 26 75 28 18 7 TARGET APPLICATION MANUFACTURER Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Wurth Electronics Pulse Engineering Pulse Engineering Pulse Engineering Pulse Engineering Pulse Engineering INPUT (V) 40 to 75 40 to 75 30 to 75 30 to 75 30 to 75 30 to 75 10 to 40 10 to 40 10 to 40 10 to 40 10 to 40 20 to 75 20 to 75 20 to 75 10 to 40 10 to 40 10 to 40 20 to 75 20 to 70 20 to 70 20 to 75 20 to 75 20 to 75 20 to 75 10 to 40 OUTPUT 12V/1A 12V/1A 12V/1A 5V/3A 5V/3A 5V/3A 5V/2.5A 5V/3A 15V/1A 24V/1.3A 15V/2A 15V/2A 24V/1.3A 12V/2.5A 12V/3.8A 15V/3A 24V/1.9A 12V/3.8A 15V/3A 24V/1.9A 12V/1A 12V/1A 12V/1A 5V/2A 10V/2.5A
*2.5k isolation, others are rated for 1.5kV isolation. TARGET APPLICATION, NOT GUARANTEED.
Turns Ratio and RMS Diode Current Note that when using an RFB/RREF resistor ratio to set output voltage, the user has relative freedom in selecting a transformer turns ratio to suit a given application. In contrast, simpler ratios of small integers (e.g., 1:1, 2:1, 3:2, etc.) can be employed to provide more freedom in setting total turns and mutual inductance. While the turns ratio can be selected to maximize output power for a given current limit, minimizing the turns ratio and increasing the current limit will often increase
efficiency and better utilize the saturation current of a given transformer. Figure 3 shows the maximum output power using three transformers with different windings ratios that have the same output inductance and peak output current, illustrating that increasing current while decreasing turns ratio can deliver more power. There are two significant constraints on the turns ratio. First, as described in the previous section on limitations to output power, the drain of the MOSFET switch will see a voltage equal to the maximum input supply plus
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the output voltage multiplied by the windings ratio plus some amount of overshoot caused by leakage inductance. Second, increasing the turns ratio will increase the peak current seen on the output diode generally increasing the RMS diode current thereby lowering the efficiency. This efficiency limitation is worse at lower output voltages when the diode forward voltage is significant compared to the output voltage. In a typical application such as the 5V, 2A output shown on the back page, the diode losses dominate all the other losses, as shown in Figure 4. To calculate RMS diode current, two equations are needed--the first for calculating duty cycle, D, and the second to calculate the RMS current of a triangle waveform: D=
25 NPS = 2:1 ILIM = 3A 20 OUTPUT POWER (W)
15 NPS = 6:1 ILIM = 1A
NPS = 3:1 ILIM = 2A
10 5
0
0
20
60 40 INPUT VOLTAGE (V)
80
100
3748 F03
( VOUT + VF(DODE) ) * NPS VIN + ( VOUT + VF(DIODE) ) * NPS
3
Figure 3. Maximum Output Power at 12V Out Using Three Transformers with Equal Peak Output Current and Secondary Inductance
100 95 EFFICIENCY LOSS (%) DOUT 90 85 80 75 70 0.2A MIN fSW * QG + IQ FET RDS(ON) TRANSFORMER I * R + LEAKAGE 2A MAX IOUT (A)
3748 F03
VIN = 12V
IDIODE(RMS) =
(ILIM * NPS )2 * (1- D)
ESTIMATED MAX LOAD EFFICIENCY (%)
For a more general analysis, Figure 5 illustrates a sweep of windings ratio on the x-axis while comparing output power and estimated efficiency for a 5V output using a 48V input. If the desired application required 20W, the maximum power curve indicates that a winding ratio of 12:1 would be sufficient at a current limit of 2A (RSENSE = 0.05), while a winding ratio of 5:1 would deliver the same power at 3A. However, when examining the corresponding efficiency at max load for those two windings ratios and current limits, the 5:1, 3A selection is clearly the superior solution with an estimated efficiency of 85% compared to 78% for the 12:1, 2A application. There are several caveats to this evaluation. First, as the diode forward voltage becomes a smaller percentage of total loss at higher output voltages (>12V) the RMS current becomes less of a concern and minimizing it will have a much smaller impact on efficiency. More significantly, if a lower turns ratio forces the use of a diode with a larger forward drop to obtain a higher reverse voltage rating, any gains from minimizing current might be lost. For low output voltages (3.3V or 5V) or high input voltages (>48V), a turns ratio greater than one can be used with multiple primary windings relative to the secondary to maximize the transformer's current gain.
Figure 4. Sources of Loss In 5V, 2A Out Typical Application
100 95 90 85 80 75 70 65 60 0 3 6 9 NPS 12 15 18
3748 F05
ILIM = 3A ILIM = 2A OUTPUT POWER
32 28 24 20 16 12 EFFICIENCY 8 4 0 MAXIMUM OUTPUT POWER (W)
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Figure 5. Estimated Efficiency and Output Power at 5VOUT from 48VIN vs Windings Ratio, NPS, at 2A and 3A Current Limits
DataSheet.in
LT3748 applications inForMation
Saturation Current As discussed earlier in the Maximum Output Power section, because the core of the transformer is being used for energy storage in a flyback, the current in the transformer windings should not exceed their rated saturation current as energy injected once the core is saturated will not be transferred to the secondary and will instead be dissipated in the core. Information on saturation current should be provided by the transformer manufacturers and Table 1 lists the saturation current of the transformers designed for use with the LT3748. Leakage Inductance and Snubbers Transformer leakage inductance (on either the primary or secondary) causes a voltage spike to appear at the primary after the MOSFET switch turns off. This spike is increasingly prominent at higher load currents where more stored energy must be dissipated. Transformer leakage inductance should be minimized. In most cases, proper selection of the external MOSFET and a well designed transformer will eliminate the need for snubber circuitry, but in some cases the optimal MOSFET may require protection from this leakage spike. An RC (resistor capacitor) snubber may be sufficient in applications where the MOSFET has significant margin beyond the predicted DC drain voltage applied in flyback while a clamp using an RCD (resistor capacitor diode) or a Zener might be a better option when using a MOSFET with very little margin for leakage inductance spiking. The recommended approach for designing an RC snubber is to measure the period of the ringing at the MOSFET drain when the MOSFET turns off without the snubber and then add capacitance--starting with something in the range of 100pF--until the period of the ringing is 1.5 to 2 times longer. The change in period will determine the value of the parasitic capacitance, from which the parasitic inductance can be determined from the initial period, as well. Similarly, initial values can be estimating using stated switch capacitance and transformer leakage inductance. Once the value of the drain node capacitance and inductance is known, a series resistor can be added to the snubber capacitance to dissipate power and critically dampen the ringing. The equation for deriving the optimal series resistance using
90 80 70 60 VDRAIN (V) 50 40 30 20 10 0 0 0.05 0.10 NO SNUBBER WITH SNUBBER CAPACITOR WITH RESISTOR AND CAPACITOR 0.15 0.20 TIME (s) 0.25 0.30
the observed periods (tPERIOD, and tPERIOD(SNUBBED)) and snubber capacitance (CSNUBBER) is below, and the resultant waveforms are shown in Figure 6. CPAR = CSNUBBER tPERIOD(SNUBBED) -1 t
2
PERIOD 2 t LPAR = PERIOD 2 CPAR * 4
RSNUBBER =
LPAR CPAR
3748 F06
Figure 6. Observed Waveforms at MOSFET Drain when Iteratively Implementing an RC Snubber
Note that energy absorbed by a snubber will be converted to heat and will not be delivered to the load. In lower power applications, the snubber may significantly reduce efficiency and in higher power applications, the snubber may need to be sized for thermal dissipation. To determine the power dissipated in the snubber resistor, it is easiest to measure the peak voltage across the snubber capacitance once the series resistance has been added and then use the following equation relating that voltage and the MOSFET switching frequency to determine the maximum power dissipation assuming the capacitor is completely discharged each cycle: PSNUBBER(MAX) = fSW * CSNUBBER * VPEAK2 Decreasing the value of the capacitor may reduce the dissipated power in the snubber at the expense of increased
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voltage on the MOSFET drain, while decreasing the value of the resistor and optionally increasing the capacitance as well will decrease the overshoot. An example of an RC snubber with minimal power dissipation but sufficient protection for the MOSFET switch is shown in the 48V, 0.5A output typical application, Figure 17. An RCD clamp, shown in Figure 7, also prevents the leakage inductance spike from exceeding the breakdown voltage of the MOSFET switch. In most applications, there will be a very fast voltage spike caused by a slow clamp diode. Once the diode clamps, the leakage inductance current is absorbed by the clamp capacitor. This period should not last longer than 200ns so as not to interfere with the output regulation. The clamp diode turns off after the leakage inductance energy is absorbed and the switch voltage is then equal to: VDS = VIN + NPS * (VOUT + VF(DIODE)) Schottky diodes are typically the best choice for use in a snubber, but some PN diodes can be used if they turn on fast enough to limit the leakage inductance spike. Figures 8 and 9 show the waveform at the drain of the MOSFET switch for the 48V output application shown in Figure 17 at maximum rated load and maximum input voltage with an RC snubber and RCD clamp, respectively. Both solutions limit the leakage spike to less than 190V, below the 200V VDS(MAX) rating of the Si7464DP MOSFET.
VIN C R LLEAK VOUT+
DRAIN VOLTAGE (V)
ring beyond that expected reverse voltage. An RC snubber or RCD clamp may be implemented to reduce the voltage spike if it is desirable to use a lower reverse voltage diode. Secondary Leakage Inductance In addition to the previously described effects of leakage inductance in general, leakage inductance on the secondary in particular exhibits an additional phenomena. It forms an inductive divider on the transformer secondary that effectively reduces the size of the primary-referred flyback pulse used for feedback. This will increase the output voltage target by a similar percentage. Note that, unlike leakage spike behavior, this phenomena is load independent. To the extent that the secondary leakage inductance is a constant percentage of mutual inductance (over manufacturing
200 180 160 140 120 100 80 60 40 20 0 0 50 100 150 200 TIME (ns) VIN = 96V VOUT = 48V IOUT = 0.5A R = 66 C = 150pF 250 300
3748 F08
Figure 8. Waveform of MOSFET Drain During Normal Operation of Figure 17 with RC Snubber (as Drawn)
200 180
GATE
3748 F07
NMOS
DRAIN VOLTAGE (V)
Leakage Inductance and Output Diode Stress The output diode may also see increased reverse voltage stresses from leakage inductance. While it nominally sees a reverse voltage of the input voltage divided by the windings ratio plus the output voltage when the MOSFET power switch turns on, the capacitance on the output diode and the leakage inductance will cause an LC tank which may
+
D VOUT-
160 140 120 100 80 60 40 20 0 0 50 100 VIN = 96V VOUT = 48V IOUT = 0.5A R = 4.99k C = TDK 0.22F 250V D = CMR1U-02M-LTC 150 200 TIME (ns) 250 300
3748 F08
Figure 7. RCD Clamp
Figure 9. Waveform of MOSFET Drain During Normal Operation of Figure 17 Using RCD Clamp with Central Semiconductor CMR1U-02M-LTC Instead of RC Snubber
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variations), this can be accommodated by adjusting the RFB /RREF resistor ratio. Winding Resistance Effects Resistance in either the primary or secondary will reduce overall efficiency (POUT /PIN). Good output voltage regulation will be maintained independent of winding resistance due to the boundary mode operation of the LT3748. Bifilar Winding A bifilar, or similar winding technique, is a good way to minimize troublesome leakage inductances. However, remember that this will also increase primary-to-secondary capacitance and limit the primary-to-secondary breakdown voltage, so, bifilar winding is not always practical. The Linear Technology Applications group is available and extremely qualified to assist in the selection and/or design of the transformer. Selecting a Current Sense Resistor The external current sense resistor allows the user to optimize the current limit behavior for the particular application under consideration. As the current sense resistor is varied from several ohms down to tens of milliohms, peak switch current goes from a fraction of an ampere to tens of amperes. Care must be taken to ensure proper circuit operation, especially with small current sense resistor values. For example, a peak MOSFET switch current of 4A requires a sense resistor of 0.025. Note that the instantaneous peak power in the sense resistor is 1W, and it must be rated accordingly. The LT3748 has only a single sense line to this resistor. Therefore, any parasitic resistance in the ground side connection of the sense resistor will increase its apparent value. In the case of a 0.025 sense resistor, 1m of parasitic resistance will cause a 4% reduction in peak switch current. Therefore, resistance of printed circuit copper traces and vias cannot necessarily be ignored. Another issue for proper operation of the current sense circuitry is avoiding prematurely tripping the SENSE threshold while slewing the MOSFET drain when the GATE pin goes high. The LT3748 does not begin to compare the SENSE pin voltage with the target threshold until the GATE pin is near its final value, or until at least 150ns has passed, whichever occurs more slowly. This should be entirely sufficient for most applications but premature tripping of the SENSE comparator may occur in cases where a MOSFET with very high QG is used with a series resistor at the GATE pin. Output Short Circuits and SENSE Pin Over Current The LT3748 has an internal threshold to detect when primary inductor current exceeds the programmed range. This can result from an inductive output short-circuit and an output voltage below zero, reflecting a voltage back to the primary side of the transformer which, in turn, causes the LT3748 to turn the external MOSFET on before the secondary current has discharged. When the voltage at the SENSE pin exceeds approximately 130mV--equivalent to 30% higher than the programmed ILIM(MAX) in the RSENSE resistor--the SS pin will be reset, stopping switching. Once the soft-start capacitor is recharged and the softstart threshold is reached, switching will resume at the minimum current limit. High Drain Capacitance and Low Current Operation When designing applications with some combination of a low current limit (ILIM < 1A), a high secondary-to-primary turns ratio (NPS << 1), multiple output windings, or very capacitive output diodes, it is important to minimize the capacitance reflected onto the primary winding and on the drain of the external MOSFET. After the MOSFET turns off during each switching cycle, the primary current charges that capacitance to slew the MOSFET drain until the secondary begins to deliver power, and if the drain node does not slew and remain above VIN within approximately 200ns once the GATE pin goes low and the MOSFET turns off, the LT3748 may detect that the current in the secondary is zero and turn the MOSFET back on prematurely, causing the LT3748 to switch continuously while delivering very little power to the output. The result will be droop of the output voltage at lighter loads and oscillation at the VC node. This problem can be prevented by maximizing NPS (minimizing ratio of secondary windings to primary windings), increasing the peak drain current (minimizing RSENSE), and minimizing the output diode and transformer capacitance.
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Soft-Start The LT3748 contains an optional soft-start function that is enabled by connecting an explicit external capacitor between the SS pin and ground. Internal circuitry prevents the control voltage at the VC pin from exceeding that on the SS pin. The soft-start function is engaged whenever power at VIN is removed, or as a result of either undervoltage lockout, overcurrent in the sense resistor or thermal (overtemperature) shutdown. The SS node is then discharged to roughly 600mV. When this condition is removed, a nominal 5A current acts to charge up the SS node towards roughly 2.2V. For example, a 0.1F soft-start capacitor will place a 0.05V/ms limit on the turn-on ramp rate at the VC node. ENABLE and Undervoltage Lockout (UVLO) A resistive divider from VIN to the EN/UVLO pin implements undervoltage lockout (UVLO). The EN/UVLO pin threshold is set at 1.223V. In addition, the EN/UVLO pin draws 2.4A when the voltage at the pin is below 1.223V. This current provides user programmable hysteresis based on the value of R1. The effective UVLO thresholds are: 1 . 223V * (R1+ R2) + 2 . 4A * R1 VIN(UVLO,RISING) = R2 VIN(UVLO,FALLING) = 1 . 223V * (R1+ R2) R2 Minimum Load Requirement The LT3748 recovers output voltage information using the flyback pulse that occurs once the external MOSFET turns off and the secondary winding conducts current. In order to regulate the output voltage, the LT3748 needs to sample the flyback pulse. The LT3748 delivers a minimum amount of energy even during light load conditions to ensure accurate output voltage information. The minimum delivery of energy creates a minimum load requirement on the output of approximately 2% of maximum load. The minimum operating frequency at minimum load is approximately 42kHz. Alternatively, a Zener diode sufficiently rated to handle the minimum load power can be used to provide a minimum load without decreasing efficiency in normal operation. In selecting a Zener diode for this purpose, the Zener voltage should be high enough that the diode does not become the load path during transient conditions but the voltage must still be low enough that the MOSFET and output voltage ratings are not exceeded when the Zener functions as the minimum load. INTVCC Pin Considerations The INTVCC pin powers the internal circuitry and gate driver of the LT3748. Three unique configurations exist for regulation of the INTVCC pin as shown in Figure 11. In the first configuration, the internal LDO drives the INTVCC pin internally from the VIN supply. In the second configuration, the VIN supply directly drives the INTVCC pin through a direct connection bypassing the internal LDO. Use this optional configuration for voltages lower than 20V. In the third configuration, an external supply or third winding drives the INTVCC pin. Use this option when a voltage supply exists lower than the input supply but higher than the regulated INTVCC voltage. Using a lower voltage supply provides a more efficient source of power for internal circuitry and reduces power dissipation in the LT3748. When calculating the minimum input voltage required for a valid INTVCC , or the power dissipated in the LT3748, it is useful to know how much current will be drawn from the INTVCC LDO during normal operation. The easiest way to calculate this current is to use the gate charge (QG) for the
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Figure 10 shows the implementation of external shutdown control while still using the UVLO function. The NMOS grounds the EN/UVLO pin when turned on, and puts the LT3748 in shutdown with a quiescent current draw of less than 1A.
VIN R1 EN/UVLO LT3748 GND
3748 F10
R2
RUN/STOP CONTROL (OPTIONAL)
Figure 10. Undervoltage Lockout (UVLO)
DataSheet.in
LT3748 applications inForMation
LT3748 VIN 5V TO 100V 3.0 2.5 INTVCC DROPOUT (V) LDO (VIN - DROPOUT) TO 7V INTVCC 2.0 INTVCC UVLO = 3.6V 1.5 1.0 0.5 0 -50 -25 VIN = 5V
IINTVCC = 20mA
LT3748
VIN
5V TO 20V
LDO
0
25 50 75 100 125 150 TEMPERATURE ( C)
3748 F12
INTVCC
OPTIONAL
Figure 12. INTVCC Current at Low VIN Can Cause the LT3748 to Stop Switching Due to INTVCC Undervoltage Lockout
LT3748
VIN
5V TO 100V
LDO
3.6V < BIAS < 20V, VIN > BIAS
and IINTVCC = 20mA might be fully functional at room temperature, but when the dropout for the same current exceeds 1.4V and trips the UVLO at higher temperatures the LT3748 will stop switching.
EXTERNAL SUPPLY OR THIRD WINDING
INTVCC
3748 F09
Overdriving INTVCC with a Third Winding The LT3748 provides excellent output voltage regulation without the need for an opto-coupler or third winding, but for some applications with input voltages greater than 20V, an additional winding may improve overall system efficiency. The third winding should be designed to output a voltage between 7.2V and 20V. For a typical 48VIN, 10W application, overdriving the INTVCC pin may improve efficiency by several percent at maximum load and as much as 30% at light loads. Loop Compensation The LT3748 is compensated using an external resistorcapacitor network on the VC pin. Typical values are in the range of RC = 50k and CC = 1nF (see the numerous schematics in the Typical Applications section for other possible values). If too large of an RC value is used, the part will be more susceptible to high frequency noise and jitter. If too small of an RC value is used, the transient performance will suffer. The value choice for CC is somewhat the inverse of the RC choice: if too small a CC value is used, the loop may be unstable and if too large a CC value is used, the transient performance will also suffer. Transient response plays an important role for any DC/DC converter.
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Figure 11. INTVCC Pin Configurations
selected MOSFET switch at the expected VIN and INTVCC voltages and multiply that charge required with each turn-on event by the maximum operating frequency. The maximum operating frequency in a given application can be approximated from the primary transformer inductance, the windings ratio (NPS), the nominal output voltage and the maximum input voltage. Unless the part is limited by minimum on- or off-times, this maximum frequency will occur when the part is regulating in boundary mode at the minimum peak switch current, and can be derived from:
fSW(MAX ) VIN(MAX ) * VOUT + VF(DIODE) * NPS L PRI * ILIM(MIN) * VOUT + VF(DIODE) * NPS + VIN(MAX )
(
(
)
)
With the maximum INTVCC current calculated, the expected dropout when VIN drops below 7V can be extracted from the curves in the Typical Performance Characteristics section. The LT3748 is tested as low as VIN = 5V but the hard limit on minimum VIN operation is the INTVCC regulator dropout and the 3.6V under voltage lockout. Figure 12 illustrates an example where operation with VIN = 5V
DataSheet.in
LT3748 applications inForMation
DESIGN EXAMPLE: 12VIN TO 5V, 2A OUT The first example is an automotive application shown on the back page of this data sheet--a nominal 12VIN, 5VOUT at 2A with an operating input voltage range of 6V to 45V with a design focus of maximizing efficiency. 1. Select Transformer Turns Ratio Transformer turns ratio will affect the requirements for the MOSFET switch VDS rating, the output diode reverse bias rating, the output power capability, and the efficiency of the overall converter. Because the output voltage is low compared to the forward drop on the output diode and the currents are high in this application, efficiency can be optimized by minimizing the RMS diode current. Although typical efficiency in a variety of applications will be 85% to 90%, due to compromises made for the wide input voltage range and due to the low output voltage in this specific application, an efficiency of 80% is assumed for calculating output power. This assumption can be revised once the application is tested. Equations for evaluating each of the important criteria are: NPS = NP/NS VDS(MAX) VIN(MAX) + VOUT * NPS VR(DIODE) VIN(MAX)/NPS + VOUT IOUT(MAX) 0.80 * (1 - D) * NPS * ILIM/2 D = (VOUT + VF(DIODE)) * NPS/(VIN + (VOUT + VF(DIODE)) * NPS) IDIODE(RMS) = (ILIM * NPS)2 * (1 - D)/3 The equation for output power can be rearranged to solve for the current limit, ILIM, which can be solved at the nominal or the minimum VIN depending on application requirements. In this application the 2A load requirement will be set at VIN = 9V to reduce operating stresses at higher input voltages. The results of the aforementioned equations in this application are found in Table 2.
NPS 0.5 1 2 3 VDS(MAX) 47.5 50 55 60 VR(DIODE) 95 50 27.5 20 D (VIN = 12V) 0.19 0.31 0.48 0.58
Evaluating the results of the table, the 1:2 turns ratio looks demanding in terms of diode reverse-voltage requirements (a diode with higher reverse bias capability generally will have a larger forward drop and therefore lower application efficiency) and primary side currents and only decreases the output diode RMS current by 13% from the 1:1 case. However, on evaluating the minimum and maximum inductance requirements in Step 3, even the 1:1 case does not allow for enough on-time from maximum VIN for the range of inductance that provides sufficient off-time. For that reason, a 2:1 turns ratio is selected, easing the requirement on the output diode reverse voltage rating in the process. 2. Calculate Sense Resistor Value The sense resistor can be calculated by the following equation: RSENSE = 100mV ILIM
The desired 6A current limit leads to an unusual value of 0.017, so the current limit is increased to use a more standard 0.016 value and ILIM of 6.25A. 3. Select a Transformer Based on Inductance and Saturation Current Requirements The transformer in this application will be selected to optimize efficiency at a 80kHz minimum switching frequency at maximum load from the nominal input voltage. In applications where transformer size is the primary requirement, reducing the current limit or increasing the switching frequency may be required. The following equations select the inductance required for a given switching frequency at max load and then verify that the inductance is large enough to satisfy the minimum on and off times of the LT3748.
Table 2. Voltage Stresses, Output Capability and Diode Current vs Turns Ratio in 12VIN to 5V, 2A Application
D (VIN = 9V) 0.23 0.38 0.55 0.65 ILIM (2A OUT AT VIN = 9V) 13 8 6 5 IDIODE(RMS) (VIN = 12V) 3.4 3.9 4.6 5.3
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DataSheet.in
LT3748 applications inForMation
LPRI VIN(MIN) * (VOUT + VF(DIODE)) * NPS/(fSW(MIN) * ILIM * ((VOUT + VF(DIODE)) * NPS + VIN(MIN))) LPRI (VOUT + VF(DIODE)) * RSENSE * 400ns * NPS/15mV LPRI VIN(MAX) * RSENSE * 200ns/15mV For this application, the primary inductance with a 2:1 transformer and a 0.016 sense resistor for an 6.25A current limit is bounded by the minimum desired switching frequency and the minimum off time requirement to be between 9.6H and 11.5H. Looking at Table 1, there are no transformers that fit that exact requirement. For the sake of prototyping, a transformer with slightly less than the desired primary inductance is selected with the PA3177NL. The application will need to be tested thoroughly for stability at higher input voltages and when the current limit is at a minimum (in the middle of the output load range). The easiest solution to ease the requirement on minimum on-time is to reduce the maximum VIN voltage although alternatively NPS could be increased at the expense of efficiency (and requiring a more thorough redesign). 4. Select a MOSFET Switch The selected 2:1 transformer requires a nominal 55V rating on the MOSFET switch, assuming no leakage inductance. However, even a small amount of leakage inductance may cause the drain to ring to double the anticipated voltage, and generally this needs to be verified in the final design. However, at currents below 10A it is fairly easy to find a MOSFET with sufficiently low RDS(ON) to be a very small contributor to maximum load efficiency losses while similarly having a low enough QG to require minimum current and minimal losses when driving the MOSFET at lighter loads. Also, while considering the efficiency gains and losses with a given MOSFET, it is important to realize that a trade-off in RDS(ON) for VDS(MAX) may backfire if a snubber needs to be added to the circuit to meet the voltage requirements and dissipates more energy than the difference in switch resistance. For that reason, a Vishay Si7738 is selected to give lots of margin with its 150V rating. The RMS current in the MOSFET can be calculated, squared and multiplied by the RDS(ON) to calculate losses and the current required to drive the FET at frequency can be determined, by the following equations: IMOSFET(RMS) = ILIM2 * D/3 IINTVCC = fSW * QG PINTVCC = IINTVCC * (VIN - VINTVCC) In this application the MOSFET RMS current at maximum load is about 2.7A, which into the 0.038 RDS(ON) will be 0.28W, or on the order of 2% loss in efficiency. Assuming that the maximum operating frequency is around four times higher than the maximum load frequency (at about a quarter the output load) and reading the approximate QG at 7V operation from the Vishay data sheet, the approximate INTVCC current is likely close to 8mA, dissipating 0.04W when the load is on the order of 2.5W, or less than 2%, and much less at maximum load. 5. Select the Output Diode The output diode reverse voltage, as calculated earlier, is the first important specification for the output diode. As with the MOSFET, choosing a diode with enough margin should preclude the use of a snubber. The second criterion is the power requirement of the diode which is more difficult to correctly ascertain--some manufacturers give direct data about power dissipation versus duty cycle, which can be used with the data from the table to determine. To avoid using a snubber, a diode with a 60V reverse-bias capability and minimal forward drop was selected--in this case, the Diodes Inc. SBR 8U60P5. In this particular application where maximizing efficiency is the goal, minimizing the maximum voltage requirement on VIN may allow the use of a diode with a lower reverse bias rating and a lower forward drop which could further increase efficiency. Alternatively, if no efficient diode is available for a particular reverse bias rating, it may be more beneficial to increase the windings ratio until a diode with low forward drop can be selected and then reevaluate whether that solution with higher RMS diode current is beneficial.
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DataSheet.in
LT3748 applications inForMation
6. Select the Feedback Resistor for Proper Output Voltage Using the iterative process laid out earlier in the Applications Information section, select the feedback resistor RFB and program the output voltage to 5V. Adjust the RTC resistor for temperature compensation of the output voltage. RREF is selected as 6.04k. 7. Select the Output Capacitor The output capacitor should be chosen to minimize the output voltage ripple while considering the increase in size and cost of a larger capacitor. The following equation calculates the output voltage ripple: VMAX = L PRI * ILIM2 2 * COUT * VOUT 9. Optimize the Compensation Network To set the compensation, the application is first configured with a 22nF capacitor and 10k resistor as a starting point. A load step is applied at both light and heavy loads at the 60V maximum input voltage and the capacitance is decreased until damping decreases to the desired limit, in this case with a compensation capacitance of 2.2nF and a response implying about 60 of phase margin. After verifying stability at the minimum input voltage, as well, the compensation capacitance is doubled for safety margin. The series resistance is varied from 5k to 50k but the optimal response is observed with 25k. For best ripple performance, select a compensation capacitor not less than 1nF and select a , compensation resistor not greater than 50k. 10. Soft-Start Capacitor and UVLO Resistor Divider A soft-start capacitor helps during the start-up of the flyback converter. Select the UVLO resistor divider for the intended input operation range. These equations are aforementioned. DESIGN EXAMPLE: 48VIN TO 12V, 2A OUT The second example is a telecom application shown on the front page of the datasheet. The focus of this application is a cheap, small and simple solution. Table 3 shows the results of the initial step for selecting the turns ratio. In this example, the output diode is a much smaller efficiency loss due to the smaller voltage drop across it in ratio to VOUT so minimizing output diode current is not as important. Of greater importance is minimizing the stresses on the MOSFET and output diode and the 4:1 case seems to be the best compromise for that to avoid using a snubber on either device.
8. Add Snubber Circuitry as Necessary With the primary components selected, the application should be constructed to evaluate ringing at the drain of the MOSFET switch and to evaluate step response to optimize the compensation network. If using an RC snubber, the equations from the Applications Information section can be used or a rough estimate of component values may come from using the published leakage inductance of the transformer and selecting a snubber capacitor ranging from 2 to 3 times larger than the published MOSFET output capacitance. In this example, at maximum load while at maximum input voltage, the drain of the MOSFET switch is probed and measured to peak at approximately 125V, well below the 150V rating of the Si7738. Similarly, the anode of the output diode is probed to look at potential ringing when the MOSFET switch turns on and a peak of 45V is measured across the diode. Therefore, no snubber circuitry is required.
Table 3. Voltage Stresses, Output Capability and Diode Current vs Turns Ratio in 48VIN to 12V, 2A Application
NPS 1 2 4 6 VDS(MAX) 84 96 120 144 VR(DIODE) 84 48 30 24 D (VIN = 48V) 0.21 0.34 0.51 0.61 D (VIN = 36V) 0.26 0.41 0.58 0.68 ILIM (2A OUT AT VIN = 36V) 6 4 3 2 IDIODE(RMS) (VIN = 48V) 3.3 3.7 4.6 5.2
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DataSheet.in
LT3748 applications inForMation
20H of primary inductance is required for minimum off-time while selecting the transformer, but in order to minimize output ripple at maximum load a 60.8H transformer is selected. To meet the saturation current (12A, peak, on the secondary windings), a Versa-Pak VP4-0047-R provides a compact and efficient solution. For the MOSFET switch, since the input voltage is so high, resistive losses on the primary side will be very low so minimizing RDS(ON) is of minimum benefit. However, since the current for the gate drive is pulled from a high VIN, minimizing both QG and operating frequency is essential unless a third winding is added. The Vishay Si7464DP with , a 200V VDS(MAX) and low gate charge, keeps the INTVCC current to just over 3mA, worst-case, which when added to quiescent current will keep power dissipation in the LT3748 to just over 1/4W at 72V VIN. The output diode only nominally has 30V of reverse bias but a B360 diode is selected to ensure enough margin that a snubber will not be required. A more expensive diode with lower forward drop might recover several percent efficiency and if high temperature operation is required a diode rated for more average current at temperature might be needed, but the B360 is small and inexpensive. The rest of the design and component selection is straightforward. Suggested Layout See Figures 13 and 14 for the DC1557A demo board layout. Note the proximity of the RREF and RFB resistors (R9, R5) to the LT3748 for optimal regulation. The location of these two resistors as close to the physical pins of the LT3748 is critical for accurate regulation. In addition, the high frequency current path from the VIN bypass capacitor (C2) through the primary-side winding, the MOSFET switch and sense resistor (R10) is a very tight loop. Similarly, the high frequency current path for the MOSFET gate switching from the INTVCC capacitor through the source of the MOSFET and sense resistor is similarly small in area. For improved regulation it is recommended that the user ensure that the high current ground is kept separate or at least physically isolated from the small-signal ground used by the other ground-referenced pins.
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DataSheet.in
LT3748 applications inForMation
Figure 13. Demo Board Topside Silkscreen
Figure 14. Demo Board Topside Metal
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DataSheet.in
LT3748 typical applications
VIN 12V TYP 10F 1F 825k EN/UVLO 150k VIN RFB RREF LT3748 TC SS VC 133k 2nF 10k 4700pF 4.7F IGBT DRIVER 3-PHASE MOTOR GND GATE SENSE INTVCC 0.0125 D3 15V 300mA C3 M1 T1 1:1:1:1:1 6H 71.5k D1 15V 300mA C1 320V IGBT DRIVER
6.04k
D2 15V 300mA
IGBT DRIVER C2
IGBT DRIVER
C1-C4: 22H 25V X7R 2 D1-D4: DIODES INC. PDS3100 M1: VISHAY Si7898DP T1: COILTRONICS VERSA-PAC VP4-0075-R
D4 15V 300mA
C4
0V
3748 F15
Figure 15. Automotive IGBT Controller Supply
DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY VIN 7V TO 15V T1 1:10:10 C1 10F C2 1F C8 0.22F 50V EN/UVLO VIN RFB RREF SS TC C7 0.1F C9 100pF LT3748 GATE SENSE VC GND R6 24.9k C4 2.2nF INTVCC 50m
3748 F16
D1 R7 600k
R1 357k R2 93.1k
R3 140k
R5 10k D3 D2
VOUT+ 300V 8mA
C5
VOUT- VOUT+ 300V 8mA
R4 6.04k M1
C6
R8 600k
VOUT-
C3 4.7F
C5, C6: 0.1F 600V 2 D1, D2: CENTRAL SEMICONDUCTOR CMR1U-06M LTC M1: FAIRCHILD FDM3622 T1: WURTH ELEKTRONIK 750311486
Figure 16. 300V Isolated Flyback Converter
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DataSheet.in
LT3748 typical applications
VIN 48V TYP T1 1:1 4.7F 0.22F 825k EN/UVLO 49.9k 66 VIN RFB RREF LT3748 TC SS VC 243k 2nF 10k 4700pF 4.7F GND GATE SENSE INTVCC 0.030
3748 F17
40H
150pF 221k
4.7F 100V 3
VOUT+ 48V 0.5A
VOUT-
6.04k M1
D1: CENTRAL SEMICONDUCTOR CMR5U-02-LTC M1: VISHAY Si7464DP T1: COILTRONICS VERSA-PAC VP4-0060-R
Figure 17. 48V, 0.5A Supply from 24V to 96V Input
100 95 90 EFFICIENCY (%) 85 80 75 70 65 60 0 0.1 0.2 0.4 0.3 OUTPUT CURRENT (A) 0.5
3748 F18
VIN = 24V
VIN = 48V VIN = 96V
Figure 18. Efficiency of 48V Supply of Figure 17
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DataSheet.in
LT3748 package Description
MS Package Varitation: MS16 (12) 16-Lead Plastic MSOP with 4 Pins Removed
(Reference LTC DWG # 05-08-1847 Rev A)
1.0 (.0394) BSC
0.889 (.035
0.127 .005)
5.23 (.206) MIN
3.20 - 3.45 (.126 - .136)
4.039 0.102 (.159 .004) (NOTE 3) 16 14 121110 9
0.305 0.038 (.0120 .0015) TYP
0.50 (.0197) BSC
4.90 0.152 (.193 .006)
0.280 0.076 (.011 .003) REF
RECOMMENDED SOLDER PAD LAYOUT
0.254 (.010)
GAUGE PLANE DETAIL "A" 0 - 6 TYP
3.00 0.102 (.118 .004) (NOTE 4)
1
0.53 0.152 (.021 .006)
DETAIL "A"
0.18 (.007)
SEATING PLANE
1.10 (.043) MAX
3 5678 1.0 (.0394) BSC
0.86 (.034) REF
MSOP (MS12) 0510 REV A NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
0.17 - 0.27 (.007 - .011) TYP
0.50 (.0197) BSC
0.1016 (.004
0.0508 .002)
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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
DataSheet.in
LT3748 typical application
5V, 2A Output from Automotive Input with Continuous Operation from 6V to 45V
VIN 12V TYP 10F 825k EN/UVLO 215k VIN RFB RREF LT3748 TC SS VC 34.8k 2nF 25k 2.2nF 4.7F GND GATE SENSE INTVCC 0.016
3748 TA02
T1 2:1 8.3H 52.3k
D1
100F 10V
D2
VOUT+ 5V, 2A 50mVP-P RIPPLE VOUT-
6.04k M1 D1: DIODES INC. SBR8U60P5 D2: DIODES INC. BZT52C5V6 M1: Si7738DP T1: PULSE PA3177NL
relateD parts
PART NUMBER LT3573 LT3574/LT3575 LT3757/LT3758 LT3957/LT3958 LT1725 LT1737
(R)
DESCRIPTION 40V Isolated Flyback Converter 40V Isolated Flyback Converters 40V/100V Flyback, Boost Controllers 40V/100V Flyback, Boost Converters 20V Isolated Flyback Controller 20V Isolated Flyback Controller
COMMENTS Monolithic No-Opto Flyback with Integrated 1.25A, 60V Switch Monolithic No-Opto Flybacks with Integrated 0.65A / 2.5A 60V Switch Universal Controllers with Small Package and Powerful Gate Drive Monolithic with Integrated 5A/3.3A Switch Controller with Load Compensation Circuitry No Opto-Isolator or Third Winding Required, Up to 50W Output VIN and VOUT Limited Only by External Components VIN and VOUT Limited Only by External Components
LTC 3803/LTC3803-3 200kHz/300kHz Flyback DC/DC Controllers LTC3803-5 LTC3805/LTC3805-5 Adjustable Frequency Flyback Controllers
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Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 FAX: (408) 434-0507
LT 0710 * PRINTED IN USA
www.linear.com
LINEAR TECHNOLOGY CORPORATION 2010

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